1. Field of the Invention
This invention relates to the boost converter, and more particularly, to the two-inductor, two-switch boost converter.
2. Description of the Prior Art
The boost converter topology has been extensively used in various ac/dc and dc/dc applications. In fact, the front end of today's ac/dc power supplies with power-factor correction (PFC) is almost exclusively implemented with the boost topology. The boost topology is also used in numerous applications with a battery-powered input to generate a high output voltage from a relative low battery voltage.
Generally, the single-inductor, single-switch boost converter topology shown in FIG. 1 and its variations exhibit a satisfactory performance in the majority of applications. Nevertheless, in a number of high-power applications the performance of the boost converter can be improved by implementing the boost converter with multiple switches and/or multiple boost inductors. Multiple-switch and/or multiple-inductor boost topologies are usually employed in high-power applications with a high input voltage, or in applications where the conversion efficiency of a single-switch boost topology is significantly degraded by the reverse-recovery losses of the boost rectifier. In addition, isolated boost topologies require multiple-switch implementations and some of them may also be implemented with multiple boost inductors.
So far, a number of isolated and non-isolated multiple-switch and/or multiple-inductor topologies have been proposed, analyzed, and evaluated. For example, FIG. 2 shows the interleaved boost topology analyzed in [1]. This topology is often used in high-power PFC applications to eliminate reverse-recovery losses of the boost rectifier by operating the two boost converters at the boundary of the continuous-conduction mode (CCM) and discontinuous-conduction mode (DCM) so that the boost switches are turned on when the current through the corresponding boost rectifier is zero. Since in high-power applications the input current (boost-inductor current) ripple of a single DCM boost converter is very high, the switching instances of the two boost switches are interleaved, i.e., phase shifted for 180.degree. degrees. With the interleaving, the input current ripple is reduced and, consequently, the size of the input filter (not shown in FIG. 2) is minimized. To achieve the operation at the CCM/DCM boundary under varying line and load-current conditions, the interleaved boost converter requires a variable switching frequency control, which is often perceived as a major drawback of the circuit. In addition, the implementation of the interleaved variable-frequency control is relatively complex.
Another multiple-switch boost converter implementation suitable for applications with high input voltage is shown in FIG. 3 [1]. The main feature of this so called "three-level" converter is that its semiconductors are subject to a voltage stress equal to a half of the output voltage. As a result, the converter can be implemented with semiconductors with a lower voltage rating, which reduces both conduction and switching losses and, consequently, improves the conversion efficiency. However, in applications with a wide input-voltage range, the boost converter in FIG. 3 requires a relatively complex control since different control strategies are required when the input voltage is below one-half of the output voltage compared to that when the input voltage is higher than one-half of the output voltage. In addition, the converter may not be able to maintain approximately equal voltage across the individual output filter caps if there is an asymmetry in the duty cycles of the two switches.
Two implementations of the isolated boost converter are shown in FIGS. 4 and 5. FIG. 4 shows the implementation with a single inductor [2], whereas FIG. 5 shows the implementation with two inductors [3], [4]. The main advantage of the two-inductor implementation is a simpler transformer design since the transformer primary has only a single winding, whereas the circuit in FIG. 4 requires two primary windings. In addition, the voltage stress of the switches in the two-inductor implementation in FIG. 5 is twice as low as the voltage stress on the switches in the single-inductor implementation in FIG. 4. Namely, the voltage stress on the primary switches in the circuit in FIG. 5 is equal to the reflected output voltage to the primary, whereas the corresponding stress in the circuit in FIG. 4 is twice the reflected output voltage. The major limitation of the two-inductor circuit is its inability to regulate the load in a wide range with a constant-frequency control.